Induction motors and universal motors are currently being used in applications requiring constant speed and low horsepower, mainly because of their competitive cost. To replace such related art motors, research has been conducted on single-phase switched reluctance motors (SRMs) over the last decade. However, prior single-phase SRM machines are not generally suitable for high performance applications due to limitations such as low output power density and only a 50% duty cycle of torque generation. They also require permanent magnets or auxiliary windings for self-starting.
Because of the limitations of single-phase SRMs, there has been more attention paid to multi-phase SRM machines (i.e., having more than one phase), especially for high torque and/or high-efficiency applications. For example, two-phase SRMs may be employed as brushless motor drives in variable-speed applications, such as for home appliances and power tools. Two-phase SRMs are particularly desirable because of their relative simplicity in design and lower costs to manufacture. Various types of two-phase SRMs are described in U.S. Pat. No. 7,015,615, by K. Ramu et al., issued Mar. 21, 2006.
FIGS 1A and 1B illustrate a related art two-phase SRM 100. SRM 100 includes a stator 110 having four stator poles 115 and a rotor 120 having two rotor poles 125. Rotor 120 is adapted to rotate around a fixed shaft 130 connected to the center of rotor 120. Each of a first pair of concentric windings 140, such as copper coils, is disposed on a respective one of diametrically opposite stator poles 115A. Windings 140 may be electrically connected in series or in parallel. Similarly, a second pair of windings 150 is disposed on a respective one of diametrically opposite stator poles 115B. Windings 150 likewise may be connected in series or in parallel. FIG. 1A shows SRM 100 in a first phase. In this first phase, a current is applied through windings 140 and the resulting magnetic force causes rotor poles 125 to align with stator poles 115A. FIG. 1B shows a second phase in which a current through windings 150 causes rotor poles 125 to align with stator poles 115B. By selectively energizing windings 140 and 150, the first and second phases of SRM 100 are activated and the rotational speed of rotor 120 can be controlled.
The phase windings of a multi-phase SRM are typically energized by a control circuit associated with the SRM. As used herein, a phase winding refers to one or more windings, such as used to activate a single phase of an SRM or other brushless machine. For example, in FIGS. 1A and 1B, each set of windings 140 and 150 may constitute a different phase winding in SRM 100. Typically, an SRM control circuit has one or more switches per phase winding, for turning on and off current flow in the winding. For example, each phase winding may be associated with one or more electrical, mechanical, or electro-mechanical switches (such as a relay), including but not limited to various types of transistor switches. Again, with reference to SRM 100 shown in FIGS. 1A and 1B, at least one switch (not shown) may be used to control the current flow through phase winding 140 and at least one different switch (not shown) may control the current flow through phase winding 150. U.S. Pat. No. 7,271,564, by K. Ramu, issued Sep. 18, 2007, at FIGS. 1-4, illustrates various examples of related art multi-switch control circuits for use with multi-phase SRM machines.
One drawback to related art multi-switch SRM control circuits is their cost. That is, each switch in the control circuit is typically associated with additional circuitry for controlling its operation. For example, each switch may be implemented by a transistor switch having associated circuitry for changing the state of the switch and may be further associated with other circuit components, such as diodes, resistors, capacitors, etc. Also, because each switch in the multi-switch circuit may be independently controlled, additional circuitry may be required to implement a switch control strategy. The added circuitry associated with each of the switches tends to significantly increase both the cost and complexity of the SRM control circuit.
To overcome the disadvantages of multi-switch control circuits, single-switch control circuits have been used with multi-phase SRM machines. Single-switch circuits typically require less circuitry, such as fewer transistor switches and diodes, than multi-switch control circuits. As a result, single-switch control circuits can reduce both the cost and complexity of an SRM. Such single-switch circuits also have the advantage that they do not require multiple control strategies for controlling multiple switches. Rather, only one switch is actively controlled to trigger multiple phases of the SRM. Various single-switch SRM control circuits are disclosed in U.S. Pat. No. 7,271,564, by K. Ramu, issued Sep. 18, 2007.
FIG. 2 illustrates a single-switch control circuit 200 that can be used in a two-phase SRM. A similar single-switch control circuit is disclosed in U.S. Pat. No. 7,271,564, by K. Ramu, issued Sep. 18, 2007, for example, in FIG. 10. Control circuit 200 includes a direct current (DC) power source 210 and control circuitry 220. DC power source 210 has an alternating current (AC) voltage supply 215, a full bridge rectifier (diodes D1, D2, D3, and D4), and a source capacitor C1. Source capacitor C1 may be polarized, so as to maintain a substantially DC (i.e., constant) voltage level between its positive terminal (i.e., positive rail) and negative terminal (also referred to as negative rail, common, or ground). Those skilled in the art will appreciate that other types of power sources that supply a substantially constant voltage level and current source for use as a DC power source alternatively could be substituted, e.g., using half-bridge rectifiers or DC voltage supplies, such as batteries.
Control circuitry 220 includes a main phase winding L1 and an auxiliary phase winding L2, both having terminals electrically connected to the positive rail of DC power source 210. The negative terminal of main phase winding L1 is electrically connected to the collector terminal of a transistor switch Q1 and to the anode terminal of a diode D5. The positive terminal of auxiliary phase winding L2 is electrically connected to a positive terminal of an auxiliary capacitor C2 and to the cathode terminal of diode D5. In this context, current enters a phase winding through its positive terminal and exits the phase winding through its negative terminal. Auxiliary capacitor C2 may be a polarized capacitor having the same polarity as source capacitor C1. The negative terminal of auxiliary capacitor C2 is electrically connected to the negative terminal of source capacitor C1.
Although phase windings L1 and L2 may be spatially separated from control circuitry 220, and in some cases may be considered to form part of the SRM motor rather than part of its control circuitry, these windings are illustrated in control circuitry 220 for purposes of discussion. In some implementations, main phase winding L1 may be used to generate the majority of torque in SRM 100 and, accordingly, may have a larger amount of copper (or other electrical conductor) and/or a greater number of turns than auxiliary phase winding L2.
When current flows through main phase winding L1, a first phase of the two-phase SRM is activated. The second phase is activated when current flows through auxiliary phase winding L2. When current flows through either of phase windings L1 or L2, thus energizing the winding, the resultant magnetic energy produces a positive or negative torque in the SRM, depending on the position of rotor 120 with respect to the energized winding. For instance, if rotor poles 125 are rotating toward the energized winding's stator poles, the change in inductance at the stator poles is positive, thus producing a positive motoring torque that is output by the SRM. On the other hand, if rotor poles 125 are moving away from the energized winding's stator poles, the inductance slope is negative and a negative, regenerative torque is produced that sends energy back to DC source capacitor C1 or C2.
In operation, transistor switch Q1 directs current through either main phase winding L1 or auxiliary phase winding L2 and, as such, selects a desired phase activation for the SRM. As shown in this exemplary embodiment, the transistor switch is implemented with an NPN bipolar junction transistor whose emitter terminal is electrically connected to the common (ground) potential and whose collector terminal is connected to main phase winding L1 and diode D5. Transistor switch Q1 is turned ON and OFF by a control signal 230 applied to its base terminal. Additional control circuitry (not shown), such as a microprocessor, a digital signal processor, an application specific integrated circuit, a field programmable gate array, etc., supplies the control signal.
When transistor switch Q1 is turned ON, the DC voltage from source capacitor C1 is applied across main phase winding L1 and transistor switch Q1, causing current to flow through main phase winding L1 and transistor switch Q1. The voltage drop across the conducting transistor switch Q1 is typically negligible compared with the DC source voltage level. While transistor switch Q1 is turned ON, any current in auxiliary phase winding L2 will rapidly decay because auxiliary capacitor C2 discharges to DC voltage source capacitor C1, thus causing the voltage at auxiliary capacitor C2 to eventually equal the voltage at source capacitor C1, resulting in zero voltage across auxiliary phase winding L2. Auxiliary capacitor C2 may have a relatively small capacitance compared with DC source capacitance C1 to ensure that it can quickly discharge to DC power source 210 and attain the DC source voltage level.
When the current through main phase winding L1 exceeds a predetermined level, or some other criteria is satisfied, control signal 230 applied to transistor switch Q1 may be adjusted to turn OFF transistor switch Q1. In this case, the current through main phase winding L1 is redirected through diode D5, which becomes forward biased when transistor switch Q1 stops conducting. The redirected current quickly charges auxiliary capacitor C2 above its residual voltage, which is equal to the DC source voltage, until the auxiliary capacitor voltage exceeds the DC source voltage and causes current to flow through auxiliary phase winding L2.
When transistor switch Q1 is turned OFF, there may exist situations where auxiliary capacitor C2 generates a current in auxiliary phase winding L2 before current has finished flowing in main phase winding L1. The current through auxiliary phase winding L2 is predominantly determined by the voltage of auxiliary capacitor C2 and its effect on the current flow through phase windings L1 and L2. In such a situation, simultaneous current flow through the main and auxiliary phase windings may reduce the net torque produced by the SRM, because auxiliary phase winding L2 may produce a negative torque at the same time that main phase winding L1 generates a positive torque (or vice versa). Thus, when transistor switch Q1 changes states from ON to OFF, there exists the possibility of a net torque loss (or switching loss) in the SRM due to simultaneous current flows in main phase L1 and auxiliary phase L2 windings.
This reduction in net torque production can become particularly apparent when transistor switch Q1 is repeatedly turned ON and OFF in accordance with a pulse-width modulation (PWM) control strategy. Specifically, transistor switch Q1 typically receives a PWM control signal 230 that periodically turns transistor switch Q1 ON and OFF throughout the entire duration of the main phase conduction period. In this context, the main phase conduction period is the period in which rotor poles 125 are rotating towards main phase winding L1 so that the change in inductance at the main phase winding is positive. Accordingly, if main phase winding L1 is energized at any time during the main phase conduction period, a positive torque will be produced.
More generally, the phase conduction period or dwell time associated with a given SRM phase is the time period in which the rotor poles are rotating so as to create a positive torque should current flow through that phase's associated phase winding. The dwell angle for a given SRM phase is the angular displacement of the rotor poles during that phase's dwell time. The dwell angle is usually equal to one half of the rotor-pole pitch, and the time required to traverse the dwell angle for a particular angular speed is the dwell time.
PWM control signal 230 comprises a pulse train that periodically turns ON and OFF transistor switch Q1 throughout the duration of the main phase conduction period. The pulse width of each pulse in PWM control signal 230 establishes the amount of time transistor switch Q1 is turned ON and, thus, the amount of time a positive torque is generated by main phase winding L1. By selecting the PWM control signal 230 frequency and its pulse width (or duty cycle), the amount of positive torque produced by main phase winding L1 can be controlled. However, the net positive torque produced by main phase winding L1 may be reduced because of negative torque that is simultaneously produced in auxiliary phase winding L2 every time PWM control signal 230 switches transistor switch Q1 from ON to OFF during the main phase conduction period. To illustrate this effect, FIGS. 3A-F illustrate the undesired reduction in net positive torque production that can result when using a related art PWM control strategy.
FIG. 3A illustrates a related art timing diagram of the main-phase inductance Lm as a function of rotor-pole position. Inductance Lm is minimized when the rotor poles are most unaligned with main phase winding L1 (e.g., FIG. 1B) and main-phase inductance Lm increases to its maximum value when the rotor poles are completely aligned with main phase winding L1 (e.g., FIG. 1A). The portion of the timing diagram in which the main phase inductance Lm increases in value corresponds to the main phase conduction period, at which time any current flow through main phase winding L1 will generate a positive torque in the SRM.
For simplicity, FIG. 3A and similar timing diagrams described hereinafter illustrate a linear change of inductance as a function of rotor position. However, those skilled in the art will understand that the change in main-phase inductance Lm may be a more complicated function of the rotor position, rotor speed, and other parameters. Also, although FIG. 3A illustrates only one period of the timing diagram, e.g., showing the main-phase inductance Lm changing from a minimum value to a maximum value back to the minimum value, those skilled in the art will understand that the illustrated timing diagram may be periodic as the rotor turns. In general, the change in main phase inductance Lm is typically periodic as the rotor poles rotate at a given angular speed between their unaligned and aligned positions.
FIG. 3B illustrates a related art timing diagram of the main phase current im that is conducted through main phase winding L1 as a result of PWM control signal 230. PWM control signal 230 repeatedly turns ON and OFF transistor switch Q1 throughout the entire main phase conduction period. And FIG. 3B also illustrates the main phase current ripple caused by the toggling of transistor switch Q1. FIG. 3D illustrates a related art timing diagram of the total amount of positive torque Tem generated as a result of main phase current im flowing through main phase winding L1 during the main phase conduction period.
Because main phase current im is commutated (i.e., transferred) from main phase winding L1 to auxiliary phase winding L2 every time transistor switch Q1 is switched from ON to OFF, an auxiliary phase current also may be generated during the main phase conduction period. FIG. 3C illustrates a related art timing diagram of an auxiliary phase current ia flowing through auxiliary phase winding L2 during the main phase conduction period. Like the main phase current ripple shown in FIG. 3B, a similar auxiliary phase current ripple may result from the switching of transistor switch Q1 by PWM control signal 230.
FIG. 3E illustrates a related art timing diagram of the auxiliary phase torque Tea generated by auxiliary-phase current ia. As rotor poles 125 rotate towards main phase winding L1, the resultant increase in main phase inductance Lm corresponds to a decrease in the inductance of auxiliary phase winding L2. That is, while the rotor poles are rotating towards main phase winding L1 (i.e., increasing the main phase inductance), the rotor poles are rotating away from auxiliary phase winding L2 (i.e., decreasing the auxiliary phase inductance). Consequently, main phase current im (FIG. 3B) generates a positive torque (FIG. 3D), whereas the auxiliary phase current ia (FIG. 3C) generates a negative torque (FIG. 3E) during the main phase conduction period. Auxiliary phase current ia also may generate a positive torque at the end of the main phase conduction period, if auxiliary phase current ia continues flowing through auxiliary phase winding L2 as rotor poles 125 begin moving away from main phase winding L1 and towards auxiliary phase winding L2.
FIG. 3F illustrates a related art timing diagram of the net torque Tec generated by the SRM during the main phase conduction period. The net torque is the sum of the torques generated by main phase winding L1 and auxiliary phase winding L2. Thus, the net positive torque Tec produced by the SRM is essentially equal to the positive torque Tem produced by main phase winding L1 reduced by the negative torque Tea simultaneously produced by auxiliary phase winding L2. To maximize the net positive torque in the SRM, it is desirable to minimize the amount of negative torque Tea produced by auxiliary phase winding L2 during the main phase conduction period.
Related art PWM control strategies not only suffer the disadvantage of decreased net positive torque production, but also may exhibit unwanted acoustic noise. Specifically, when main phase winding L1 is producing a positive torque and auxiliary phase winding L2 is simultaneously producing a negative torque (e.g., prior art FIGS. 3A-F), the SRM may experience high audible noise apart from its switching losses. This undesired acoustic noise can significantly decrease the commercial attractiveness of single-switch control circuits for use in many SRM-based consumer products, such as household appliances and hand-held power tools.
All reference material cited herein is hereby incorporated into this disclosure by reference.